Receiver with a signal path

ABSTRACT

The invention relates to a receiver ( 1, 49, 51, 52, 54 ) for receiving RF signals. The known receiver comprises a phase-locked loop which is controlled by the stereo pilot. Because of unwanted frequency changes, a sampling rate converter precedes the stereo decoder. Filtering operations within a complex range can be performed by means of the invention.

The invention relates to a receiver with a signal path comprising thefollowing elements: a tuning arrangement, a demodulator circuit forsupplying a stereo multiplex signal with a baseband stereo sum signal(L+R), a 19 kHz stereo pilot and a stereo difference signal (L−R)double-sideband amplitude-modulated on a blanked 38 kHz subcarrier, asampling arrangement for converting an analog signal into atime-discrete signal, and a stereo decoder with a filter and aphase-locked loop comprising an oscillator.

Such a receiver is known from EP 0512606 B1. In the UHF range of 88-108MHz, RF signals are transmitted as frequency-modulated signals. Moststations transmit a stereo signal. After demodulation of the RFfrequency-modulated signal, a stereo multiplex signal with a basebandstereo sum signal (L+R) in a 15 kHz range and a stereo difference signal(L−R) which is double-sideband amplitude-modulated on a blankedsubcarrier of 38 kHz is obtained. The sum signal (L+R) is also referredto as mono signal. A demodulation of the stereo difference signal (L−R)requires a receiver with a large number of circuit components. Thereceiver includes a phase-locked loop which is controlled by the stereopilot. When the frequency of the transmitter changes, the stereo pilotalso changes. The demodulator in the receiver is readjusted. Because ofthese unwanted frequency changes, a sampling rate converter, referred toas SRC for short, precedes the stereo decoder. A second sampling rateconverter follows the stereo decoder. These converters are elaborate.

It is therefore an object of the invention to provide a simple stereodecoder.

This object is solved by the characteristic features defined in claim 1.According to the invention, filter operations can be performed in acomplex range. Frequency response edges are in a complex range around 0Hz. A multiplication, performed within a period of time, of a real inputsignal with a cosine wave yields a shift towards two sides within thefrequency range, i.e. a modulation around the carrier frequency +/−φ:Y(e ^(iθ))=(X(e ^(i(θ−φ))+X(e ^(i(θ+φ)))/2

A modulation by means of a cosine wave having a carrier frequency φproduces an output signal in which the interesting part is supplementedby an unwanted part of the input spectrum around +/−2φ. This can beprevented by means of a prefilter which suppresses the unwanted part inthe spectrum around +/−2φ. The same applies to a modulation with a sinewave.

A multiplication of a real or complex signal by means of a complexexponent e^(iθn), i.e. with an imaginary exponent, leads to a shift toonly one side in the frequency range so that no prefilter is used.Y(e ^(iθ))=(X(e ^(i(θ−φ)))

In the stereo decoder, complex modulations are realized by means of thesignals cos(nφ) and sin(nφ) supplied by the oscillator. Thenon-recursive half band filters, i.e. the finite impulse responsefilters, referred to as FIR filters for short, have the property of aπ/2 phase shift. This π/2 phase shift is also referred to as phasequadrature or as quadratic mirroring. The term quadratic mirroringindicates that the transfer function H(f) of this type of filter can bemirrored by a quarter of the sampling frequency (Fs/4) in accordancewith the following equation.|H(Fs/4−f)|+|H(Fs/4+f)|=1

The term half band refers to a second property of FIR filters, namely tothe fact that these filters serve for a reduction and/or aninterpolation. The FIR filters have the interesting property that halfof the coefficients is zero. When used for reduction, this means indigital techniques that every second value in a table is removed. Forinterpolation, this means that a second value, namely the precedingvalue, is inserted behind each value in the table. A twofold reductionis also referred to as down-sampling by 2.

The third interesting property of the FIR filters is that the delay isan integral multiple of the sampling when the length is chosen to beodd. When these FIR filters are used in connection with complexmodulations, only simple delay members are to be inserted so that thecomplex modulations in the stereo decoder are in phase at differenttimes. The transfer functions of the FIR filters used in the stereodecoder for complex signals are shifted by a quarter of the samplingfrequency in the frequency range so that the transition bands,hereinafter also referred to as slopes, are centered around thefrequency of 0 Hz, i.e. around f₀=0 and overlap with the L+R and L−Rspectra which can also be centered around f₀=0 when these filters areused. The value f₀=0 is also referred to as DC by analogy with directcurrent, which has the zero frequency at the applied voltage. Because ofthe mirroring property, the L+R and L−R signal can be retrieved byconnecting the real parts of the signals.

The shift of the transfer function of a FIR filter in the frequencyrange by a quarter of the sampling frequency means that the coefficientsof the real FIR filters are modified in the following way:h[n]→>h[n]e^(inπ/)2

This modification of the coefficients has no further consequences forrealizing the FIR filters.

These three properties of the FIR filters in combination with complexmodulations are the key to an elegant solution for the stereo decoder.

These and other aspects of the invention are apparent from and will beelucidated with reference to the embodiments described hereinafter.

IN THE DRAWINGS

FIG. 1 is a block diagram of a receiver including a stereo decoder,

FIG. 2 shows a first frequency spectrum at the input of the stereodecoder,

FIG. 3 shows the first spectrum and a frequency response of a first halfband, or FIR, filter,

FIG. 4 shows a second spectrum at the output of the first FIR filter,

FIG. 5 shows a third spectrum at the output of a first modulator,

FIG. 6 shows the third spectrum and a frequency response of a second FIRfilter,

FIG. 7 shows a fourth spectrum at the output of the second FIR filter,

FIG. 8 shows a fifth spectrum at the output of a second modulator,

FIG. 9 shows the fifth spectrum and two further frequency responses of asymmetrical FIR high-pass and low-pass filter,

FIG. 10 shows a sixth spectrum at a first output of the symmetrical FIRhigh-pass and low-pass filter,

FIG. 11 shows a seventh spectrum at the second output of the symmetricalFIR high-pass and low-pass filter,

FIG. 12 shows a pilot at an output of an elliptic filter,

FIG. 13 shows an eighth spectrum with a complex L+R signal at the outputof a third modulator,

FIG. 14 shows a ninth spectrum with a complex L−R signal at the outputof a fourth modulator,

FIG. 15 shows a tenth spectrum of a real L+R signal at the output of afirst converter,

FIG. 16 shows an eleventh spectrum of a real L−R signal at the output ofa second converter,

FIG. 17 is a block diagram of a phase-locked loop, and

FIG. 18 is a block diagram of an oscillator.

FIG. 1 shows a stereo decoder 1 with a finite impulse response, or FIR,filter 2, a complex modulator 3, a second FIR filter 4, a second complexmodulator 5, a down-sampling-by-2 filter 6, a circuit 7 with two FIRfilters 8 and 9, a third and a fourth modulator 10 and 11, two furtherdown-sampling-by-2 filters 12 and 13, two converters 14 and 15, anelliptic low-pass filter 16, a control path 17, a double interpolationfilter 18, an oscillator 19, a delay member 20, a fifthdown-sampling-by-2 filter 21, a second delay member 22, a sixthdown-sampling-by-2 filter 23 and a third delay member 24. Input signalsare applied to the FIR filter 2 via an electrically conductiveconnection 25 in the stereo decoder 1. Two further electricallyconductive connections 26 lead from the FIR filter 2 to the modulator 3and apply signals from the FIR filter 2 to the modulator 3. Signals fromthe modulator 3 are applied to the second FIR filter 4 via twoelectrically conductive signal connections 27. Signals are furtherapplied to outputs 37 and 38 via further signal connections 27 to 36 andvia the FIR filter 4, the modulator 5, the FIR filters 8 and 9, themodulators 10 and 11, the down-sampling-by-2 filters 12 and 13, and theconverters 14 and 15. The connections 26 to 36 are two parallelconnections each transmitting a signal.

The oscillator 19 is a discrete controlled oscillator, referred to asDCO for short. The DCO 19 has three outputs with two electricallyconductive signal connections 39 to 41 which lead to the complexmodulator 3, via the delay member 20 and a further connection 42 to themodulator 5 and via the down-sampling-by-2 filter 21 and the seconddelay member 22 and further connections 43 and 44 to the modulator 10,via the FIR filter 4, the down-sampling-by-2 filter 23 and the thirddelay member 24 and further connections 45, 46 and 47 to the modulator11. The DCO 19 generates a cosine signal on one signal connection of anoutput and a sine signal on the other signal connection. The signalshave a frequency of 38 kHz on the connection 39, a frequency of +19 kHzon the connections 40, 45, 46 and 47, and a frequency of −19 kHz on theconnections 41, 42, 43 and 44.

A tuning arrangement 49 with an antenna 50, a frequency modulator 51 andan A/D converter 52 are arranged at an input 48 of the stereo decoder 1.The converter samples the time-division multiplex signal with a samplingrate Fs of 4×44.1 kHz. The tuning arrangement 49 is controlled via aconnection 53. Arranged at the outputs 37 and 38 of the stereo decoder 1is a converter 54 which generates a left and a right stereo signal fromthe mono signal L+R and the difference signal L−R, which stereo signalsare reproduced as acoustic signals by loudspeakers 55 and 56. The stereodecoder 1, the tuning arrangement 49, the frequency modulator 51, theA/D converter 52 and the converter 54 constitute a receiver.

The FIR filters 2, 4, 7, 8 and 9 in combination with complex modulationsare the key to an elegant solution for the stereo decoder 1 whosefunction will now be elucidated with reference to FIGS. 2 to 15.

FIG. 2 shows a spectrum of a multiplex signal applied to the stereodecoder 1, which signal is present on the connection 25 and is sampledat a sampling rate Fs of 4×44.1 kHz. The spectrum is shown without RDS,ARI and SCA signal. Starting from zero, the baseband stereo sum signalL+R with the baseband 57, the pilot 58 at 19 kHz and subsequently thestereo difference signal L−R with the two sidebands 59 and 60double-sideband amplitude-modulated on a 38 kHz subcarrier extend in theright half of the spectrum. Because of the symmetry property within thefrequency range, the bands and the pilot 57-60 are mirrored around zeroand occur in a side-inverted form in the left half of the spectrum asbands and pilots 61, 62, 63 and 64.

FIG. 3 shows a frequency response 65 of the symmetrical FIR low-passfilter 2 which, viewed from the zero-crossing, is shifted to the rightby Fs/4, i.e. by 44.1 kHz. The L+R signal is thus in the transmissionband 66, which is hereinafter also referred to as slope. The filter 2 iscomplex, operates in a complex manner and also supplies a complex outputsignal.

FIG. 4 shows a spectrum of the complex output signal after filtering ofthe filter 2. Since the L+R signal is filtered with slope values withinthe slope 66, reduced values, dependent on the relevant slope value, areobtained for the L+R signal. Sidebands 67 and 68 of the L+R signal arereduced. The complex output signal of the filter 2 is present on theconnection 26.

FIG. 5 shows a spectrum after the modulation by the modulator 3. Thesignal is complex-modulated at −38 kHz in the modulator 3, i.e. thespectrum is shifted to the left by −38 kHz. The L−R signal of thespectrum is thus centered around zero, i.e. around DC. The zero is nowbetween the two sidebands 59 and 60 of the L−R signal. The output signalof the modulator 3 is supplied on the connection 27.

FIG. 6 shows the centered L−R signal which is now applied to thesymmetrical FIR filter 4. The filter is shifted to the left by Fs/4,i.e. by 44.1 kHz. A filtering with the symmetrical FIR high-pass filter,shifted to the right by Fs/4, is also possible. The L−R signal, i.e. thetwo sidebands of the L−R signal, are thus situated in a secondtransition band 69, hereinafter also referred to as slope, of a secondfrequency response 70.

FIG. 7 shows a spectrum after the filtering by means of the filter 4.Since the stereo difference signal L−R is filtered with slope valueswithin the slope 69, reduced values, dependent on the relevant slopevalue, are obtained for the L−R signal. The associated signal withreduced sidebands 71 and 72 is supplied on the connection 28 and appliedto the modulator 5.

FIG. 8 shows the spectrum complex-modulated at 19 kHz in the modulator 5and shifted to the right by 19 kHz. When the frequencies of the complexmodulation are exact multiples of the original pilot frequency, thepilot is now situated at the zero-crossing. The signal is down-sampledby 2 in the down-sampling-by-2 filter 6. From the connection 30, thecomplex signal is passed through two different branches. In one branch,the signal is applied to the filter circuit 7 for the purpose of audioprocessing and in the other branch it is applied to an elliptic filter16, i.e. a bandpass filter having a small bandwidth for extraction ofthe pilots 58 and 62. The pilot 58, which is now near DC, is used forcontrolling the DCO 19 which controls the complex modulations.

FIG. 9 shows the signal in the filter circuit 7. The FIR filter 8 with afrequency response 73 is shown in the left-hand part and the FIR filter9 with a frequency response 74 is shown in the right-hand part. Thefilter circuit is a symmetrical FIR high-pass and low-pass filter whichis shifted to the left by (Fs/2)/4=22.05 kHz so that the L+R and the L−Rsignal are separated.

FIG. 10 shows a spectrum of an output signal as supplied by the FIRlow-pass filter 8 on the connection 32. The signal is the L+R monosignal complex-filtered with the slope 66, with the two reducedsidebands 67 and 68.

FIG. 11 shows a spectrum of an output signal as supplied by the FIRfilter 9 on the connection 31. The signal is the L−R stereo differencesignal complex-filtered with the slope 69, with the two reducedsidebands 71 and 72.

FIG. 12 shows a spectrum after the low-pass filter 16. The pilot 58 isat DC.

FIG. 13 shows the spectrum of the L+R mono signal after the modulator10. In the modulator 10, the signal is modulated with 19 kHz, i.e.shifted to the right by 19 kHz, so that the two reduced sidebands 67 and68 of the spectrum are DC-centered.

FIG. 14 shows the spectrum of the L−R difference signal after themodulator 11. In the modulator 11, the L−R signal is modulated with −19kHz, i.e. shifted to the left by −19 kHz so that the two reducedsidebands 71 and 72 of the spectrum are DC-centered.

FIG. 15 shows a spectrum of the L−R signal with original sidebands 75and 76 after the converter 14. The converter 14 filters the real partsfrom the complex L+R signal, thus obtaining the original L+R signal.

FIG. 16 shows a spectrum of the L−R signal with original sidebands 77and 78 after the converter 15. The converter 15 filters the real partsfrom the complex L−R signal, thus obtaining the original L−R signal.

FIG. 17 shows a phase-locked loop, or PLL, 80 with the modulator 3, theFIR filter 4, the second modulator 5, the down-sampling-by-2 filter 6,the elliptic low-pass filter 16, the control path 17, the interpolationfilter 18, the DCO 19, and the delay member 20. The control path 17comprises an amplifier 81 with a coefficient a, a delay member 82 and asecond amplifier 83 with a coefficient b in a forward control 84, and adelay member 85 in a feedback control 86, as well as two adders 87 and88. The PLL 80 operates as follows.

The original L−R signal can only be regained exactly and in phase withthe L+R signal when the DCO 19 is clocked with the pilot in frequencyand phase synchronism. This means that the complex signal has only a DCpart after the elliptic low-pass filter 16, or the imaginary part of thesignal is zero. Deviations from zero are used to control the DCO 19 inphase synchronism with the pilot by means of the PLL 80.

When the offset, starting from the initial phase and frequencydeviation, is to be set to zero, a proportional and integrating controlpath 16 is necessary so that the input signal, which is step-shaped bothin phase and in frequency, is synchronous with zero in the offset.

Only the imaginary part after the complex modulation, i.e. actually onlythe phase recognition is utilized in the feedback loop of the PLL and isused for controlling the DCO 19.

The properties of the transient response such as response time andattenuation are adjustable by adjustment of the multiplicationcoefficients a and b of the amplifiers 81 and 83 in the control path 17.

The input signal of the oscillator 19 is a correction of the mismatchingbetween the phase of the pilot and the output signal of the DCO 19.

FIG. 18 shows the DCO 19 with four operational amplifiers 90, 91, 92 and93, two delay members 94 and 95 and two adders 96 and 97. The complexoscillator 19 generates a cosine signal at a first output 98 and a sinesignal at a second output 99. Coefficients c of the operationalamplifiers 90 and 92, as well as coefficients s and −s of theoperational amplifiers 91 and 93 can be calculated as follows:c=cos(2πθ/Fs)s=sin(2πθ/Fs)

The original values in the delay circuits 94 and 95 should be set to 0and 1. The output signal of the control path, being a correction of themismatching, is used to adapt the coefficients c and s by linear Taylorsequences, in which En is the output signal of the control path 17,which controls the DCO 19:c=cos(2πθ/Fs)−sin(2πθ/Fs)*Σεns=sin(2πθ/Fs)+cos(2πθ/Fs)*Σεn

The complex oscillator 19 with the oscillation frequency Θ may be formedin software as a limit-stable oscillating filter.

List of Reference Numerals:

-   1 stereo decoder-   2 FIR filter-   3 complex modulator-   4 second FIR filter-   5 complex modulator-   6 down-sampling-by-2 filter-   7 filter circuit-   8, 9 FIR filter-   10, 11 complex modulator-   12, 13 down-sampling-by-2 filter-   14, 15 converter-   16 low-pass filter-   17 control path-   18 interpolation filter-   19 oscillator-   20 delay member-   21 down-sampling-by-2 filter-   22 second delay member-   23 down-sampling-by-2 filter-   24 third delay member-   25, 26, 27,-   28, 29, 30, 31,-   32, 33, 34,-   35, 36 signal connections-   37, 38 output-   39, 40, 41 signal connections-   42, 43, 44,-   45, 46, 47 connections-   48 input-   49 tuning arrangement-   50 antenna-   51 frequency demodulator-   52 A/D converter-   53 connection-   54 converter-   55, 56 loudspeaker-   57 L+R signal-   58 pilot-   59 L−R signal first sideband-   60 L−R signal second sideband-   61 L+R signal side-inverted-   62 pilot, side-inverted-   63 L−R signal first band, side-inverted-   64 L−R signal second band, side-inverted-   65 frequency response-   66 slope-   67 L+R sideband, reduced-   68 second L+R sideband, reduced-   69 second slope-   70 second frequency response-   71 L−R sideband, reduced-   72 second L−R sideband, reduced-   73, 74 frequency response-   75, 76 real L+R sideband-   77, 78 real L−R sideband-   79-   80 phase-locked loop-   81 amplifier-   82 delay member-   83 amplifier-   84 forward control-   85 delay member-   86 feedback-   87, 88 adder-   89-   90, 91,-   92, 93 operational amplifier-   94, 95 delay member-   96, 97 adder-   98, 99 output

1. A receiver (1, 49, 51, 52, 54) with a signal path comprising thefollowing elements: a tuning arrangement (49), a demodulator circuit(51) for supplying a stereo multiplex signal with a baseband stereo sumsignal (L+R), a 19 kHz stereo pilot and a stereo difference signal (L−R)double-sideband amplitude-modulated on a blanked 38 kHz subcarrier, asampling arrangement (52) for converting an analog signal into atime-discrete signal, and a stereo decoder (1) with a filter (2, 4, 7,8, 9) and a phase-locked loop (80) comprising an oscillator (19),characterized in that filter operations can be performed in a complexrange.
 2. A receiver as claimed in claim 1, characterized in that thefilter (2, 4, 7, 8, 9) is complex.
 3. A receiver as claimed in claim 1,characterized in that the complex filter (2, 4, 7, 8, 9) is a finiteimpulse response filter (2, 4, 7, 8, 9).
 4. A receiver as claimed inclaim 1, characterized in that the oscillator (19) isdiscrete-controlled.
 5. A receiver as claimed in claim 1, characterizedin that the oscillator (19) supplies a complex signal.
 6. A receiver asclaimed in claim 1, characterized in that the oscillator (19) supplies acosine signal and a sine signal.
 7. A receiver as claimed in claim 1,characterized in that the oscillator (19) comprises a limit-stableoscillating filter.
 8. A receiver as claimed in claim 1, characterizedin that the oscillator (19) controls a modulator (3, 5, 10, 11).
 9. Areceiver as claimed in claim 8, characterized in that the modulator (3,5, 10, 11) comprises a multiplying member.
 10. A receiver as claimed inclaim 1, characterized in that the sampling arrangement (52) operates ata fixed clock.
 11. A receiver as claimed in claim 10, characterized inthat the fixed clock is between 4×20 kHz and 4×80 kHz, advantageouslybetween 4×32 kHz and 4×64 kHz, particularly at 4×44.1 kHz.
 12. Areceiver as claimed in claim 1, characterized in that the stereo pilotis filtered with an elliptic filter (16) having a frequency responsearound 0 Hz.
 13. A receiver as claimed in claim 1, characterized in thatthe stereo decoder (1) comprises a converter (14, 15) which convertscomplex signals to real signals.
 14. A receiver as claimed in claim 1,characterized in that the phase-locked loop (80) comprises a controlpath (17) with an amplifier (81, 83).
 15. A method of decoding atime-discrete stereo multiplex signal with a baseband stereo sum signal(L+R), a 19 kHz stereo pilot and a stereo difference signal (L−R)double-sideband amplitude-modulated on a blanked 38 kHz subcarrier in adecoder of a receiver, characterized by the steps of filtering thestereo multiplex signal by means of a filter, in which one of the twostereo signals (L+R, L−R) is complex-filtered by means of a slope,complex-modulating the filtered signal by means of a modulator,filtering the modulated signal by means of a filter, in which the otherone of the two stereo signals (L+R, L−R) is complex-filtered by means ofa slope, complex-modulating the signals, separating the baseband stereosum signal (L+R) and the stereo difference signal (L−R), modulating theL−R and the L+R signal, and converting the signals from complex signalsto real signals.
 16. A method as claimed in claim 15, characterized inthat the modulated signal is down-sampled by 2 after the secondmodulation.
 17. A method as claimed in claim 15, characterized in thatthe signal is down-sampled by 2 after the third modulation.
 18. A methodas claimed in claim 15, characterized in that the real signals areseparated into a left and a right stereo signal.